Variable transconductance amplifier

ABSTRACT

A variable transconductance amplifier including a variable attenuator stage coupled to a transconductance stage. The variable attenuator includes first and second differential to single-ended transconductance stages, each biased by a current device. The variable attenuator receives a differential input voltage signal and develops a current signal. At least one reactive element is coupled between the pair of differential to single-ended transconductance stages. The transconductance stage includes first and second differential pairs each having first and second control terminals and first and second output terminals. The first and second differential pairs are coupled to the first and second differential to single-ended transconductance stages, respectively, of the variable attenuator. The output terminals of the first and second differential pair are cross-coupled to develop a differential output current signal. The stages include electronically controllable current devices so that the overall transconductance decreases when the input signal increases without distorting the output signal.

CROSS-REFERENCE TO RELATED APPLICATION(S)

The present application is based on U.S. Provisional Patent Applicationentitled “Variable Transconductance Amplifier”, Ser. No. 60/257,763,filed Dec. 21, 2000, which is hereby incorporated by reference in itsentirety. The present application is related to U.S. Patent Applicationentitled “A Calibrated DC Compensation System For A WirelessCommunication Device Configured In A Zero Intermediate FrequencyArchitecture”, Ser. No. 09/677,975, filed Oct. 2, 2000, which is herebyincorporated by reference in its entirety.

FIELD OF THE INVENTION

The present invention relates to filters, converters and amplifiers, andmore particularly to a variable transconductance amplifier with improvedlinearity and that maximizes input voltage utilization.

DESCRIPTION OF RELATED ART

The bipolar junction transistor (BJT) differential pair is one of themost important integrated circuit building blocks. It can be used as atransconductance stage for variable gain amplifiers and g_(m)-C filters.The differential output current I_(D) is related to the emitter biascurrent I_(E) and the input differential voltage V_(Dif) in accordancewith the following equation 1:

I _(D) ˜I _(E) tan h[V _(Dif)/2V _(T)]˜(I _(E)/2V _(T))V _(Dif)  (EQ 1)

where “˜” denotes “approximately equal” and where V_(T) is a thermalcoefficient voltage (the voltage equivalent of temperature, whereV_(T)=kT/q, where “k” is the Boltzmann constant in joules per degreeKelvin, T is the temperature in degrees Kelvin [absolute scale], and “q”is the magnitude of the charge of an electron), and when V_(Dif)<V_(T).

The transconductance (g_(m)), which is the ratio of output current toinput voltage, can be scaled by changing the emitter bias current. Theonly major problem with this stage is the maximum useful input voltage.Significant distortion occurs when the peak input voltage is in excessof about 2V_(T). Linearity can be improved by adding resistive orinductive emitter degeneration, but then the transconductance cannot bechanged electronically.

Prior art approaches to increasing the allowable input voltage (overthat of a single differential pair) include dividing the input voltageacross series-connected differential pairs or using an attenuator infront of the differential pair(s). The attenuator usually has low inputimpedance and does not lend itself to electronic gain control. Theseries-connected differential pairs (multi-tanh) becomes impractical forlarge input voltages, where “tanh” refers to the hyperbolic tangentfunction.

SUMMARY OF THE INVENTION

A variable transconductance amplifier according to the present inventionincludes a variable attenuator stage coupled to a transconductancestage. The variable transconductance amplifier allows the overalltransconductance to be decreased when the input signal is increasedwithout distorting the output signal. Embodiments of the variableattenuator stage described herein exhibit a relatively high inputimpedance and are electronically controllable. Embodiments of thetransconductance stage also have electronically controllable gain. Inthis manner, electronic gain control can be applied to either or bothstages so that attenuation may be increased in the presence ofincreasing input voltage to maintain linearity. In various embodimentsdescribed herein, electronic gain control is achieved usingelectronically controllable current devices, such as current sinks orcurrent sources, that may be controlled by external circuitry as isknown to those skilled in the electronic arts.

In a particular embodiment, the variable attenuator includes first andsecond differential to single-ended transconductance stages, where eachstage is biased by a respective one of first and second current devices.Also, each stage has a first control terminal, so that both stagescollectively receive a differential input voltage signal. Each stagefurther has a second control terminal that develops a current signal. Atleast one reactive element or a reactive circuit is coupled between thesecond control terminals of the pair of differential to single-endedtransconductance stages, so that the developed current signal flowsthrough this element or circuit. In this embodiment, thetransconductance stage includes a first differential pair having firstand second control terminals and first and second output terminals,where the first and second control terminals of the first differentialpair are coupled to the first and second control terminals,respectively, of the first differential to single-ended transconductancestage of the variable attenuator. The transconductance stage furtherincludes a second differential pair having first and second controlterminals and first and second output terminals, where the first andsecond control terminals of the second differential pair are coupled tothe first and second terminals, respectively, of the second differentialto single-ended transconductance stage of the variable attenuator. Thefirst output terminal of the first differential pair is coupled to thesecond output terminal of the second differential pair and forms a firstpolarity of a differential output current signal. Also, the first outputterminal of the second differential pair is coupled to the second outputterminal of the first differential pair and forms a second polarity ofthe differential output current signal.

The particular configuration of the reactive element depends upon thespecific application in which the variable transconductance amplifier isemployed. In one embodiment, a single resistor may be used. Inalternative embodiments, an inductor, a capacitor, or any combination ofsuch components may be employed as appropriate for the specificapplication and signal frequency. For example, the combination ofcomponents may have a characteristic frequency or frequency responsedesigned for specific applications.

The first and second current devices may be electronically controllableto adjust the gain of the variable attenuator and to adjust the maximumallowable input voltage of the differential input voltage whilemaintaining linearity. In a similar manner, the first and seconddifferential pairs may each be biased by at least one additional currentdevice that is electronically controllable to adjust thetransconductance between the differential input voltage signal and thedifferential output current signal. In one embodiment, the first andsecond differential pairs of the transconductor stage may each have abias terminal, where the bias terminals of the first and seconddifferential pairs may be coupled together. Also, a third current devicemay be coupled to the common bias terminals of the first and seconddifferential pairs. This third current device may be electronicallycontrollable as the first and second current devices.

In one alternative embodiment of the variable attenuator, eachdifferential to single-ended transconductance stage includes adifferential pair of transistors including a diode-coupled transistor, acurrent sink and a current mirror coupled to the differential pair. Inanother alternative embodiment of the variable attenuator, eachdifferential to single-ended transconductance stage includes adifferential pair of transistors including a first transistor coupled toa voltage supply signal and a diode-coupled transistor. The first andsecond current devices are each current sinks coupled to a respectiveone of the differential pairs. Also, for each differential pair, acurrent source is coupled to the diode-coupled transistor, where thecurrent source sources a current level that is proportional to thecurrent developed by the corresponding current sink of the differentialpair.

In yet another embodiment, first and second current sources are providedfor each differential pair, where each is coupled to source current to arespective one of the transistors of the differential pair. In thislatter configuration, the current sources each source a current that isproportional to the current developed by a corresponding current sink ofthe differential pair. In yet another alternative embodiment, thedifferential pairs of transistors of the first and second differentialto single-ended transconductance stages are cross-coupled. Capacitorsmay be added in the cross-coupled configuration to facilitate operationat higher frequencies.

Many variations are possible for each of the variable attenuator andtransconductance stages of the variable transconductance amplifier. Forexample, different types of transistors may be used, such as NPN or PNPbipolar junction transistors, metal-oxide semiconductor, field-effecttransistors (MOSFETs), etc. The current mirrors, for example, may beconfigured using common base PNP bipolar junction transistors as knownto those skilled in the art. Current mirrors, if used, may have a 1:1current ration or any other desirable current ratio. The relativecurrent levels between corresponding current sources and current sinksof each portion of each stage may be altered if desired for certainconfigurations. The transistors of each transistor pair or differentialpair configuration may have different sizes, emitter areas, currentcapacities, etc. Level shifting elements may be added if desired, suchas for configurations required to handle large input voltages that mightotherwise drive certain transistors into saturation. For example, diodesmay be connected in series with the bases of selected bipolar junctiontransistors. For high frequency operation, capacitors may be coupledbetween the each of the first and second differential to single-endedtransconductance stages of the variable attenuator and the reactiveelement.

BRIEF DESCRIPTION OF THE DRAWINGS

A better understanding of the present invention can be obtained when thefollowing detailed description of embodiments of the invention isconsidered in conjunction with the following drawing, in which:

FIG. 1 is a block diagram of an exemplary variable transconductanceamplifier in accordance with an embodiment of the present inventionincluding a variable attenuator coupled to a transconductance stage.

FIG. 2 is a schematic diagram of an exemplary variable attenuator thatmay be used as the variable attenuator of FIG. 1.

FIG. 3 is a schematic diagram of an exemplary transconductance stagethat may be used as the transconductance stage of FIG. 1.

FIG. 4 is a schematic diagram of another exemplary variable attenuatorthat may be used as the variable attenuator of FIG. 1.

FIG. 5 is a schematic diagram of another exemplary variable attenuatorthat may be used as the variable attenuator of FIG. 1.

FIG. 6 is a schematic diagram of another exemplary variable attenuatorthat may be used as the variable attenuator of FIG. 1.

FIG. 7 is a schematic diagram of an exemplary transconductance stagethat may be used as the transconductance stage of FIG. 1.

DETAILED DESCRIPTION OF EMBODIMENT(S) OF THE INVENTION

FIG. 1 is a block diagram of an exemplary variable transconductanceamplifier 100 in accordance with an embodiment of the present invention.The transconductance amplifier 100 includes a first stage variableattenuator 101 that reduces the voltage applied to a secondtransconductance stage 103. Electronic gain control is applied to eitheror both the variable attenuator 101 and the transconductance stage 103.The attenuation may be increased in the presence of increasing inputvoltage to maintain linearity. The variable attenuator 101 is referencedbetween a voltage supply signal V_(Supply) and a common referencesignal, such as ground, and receives a differential input voltage V_(IN)having differential polarity signals V_(IN+) and V_(IN−). It isappreciated that any suitable reference signal is contemplated,including ground. The variable attenuator 101 establishes four voltagesignals V_(A), V_(B), V_(C) and V_(D) (having voltages of V_(A), V_(B),V_(C) and V_(D), respectively) to the transconductance stage 103, whichgenerates a differential output current I_(OUT) having differentialpolarity output current signals I_(OUT+) and I_(OUT−).

FIG. 2 is a schematic diagram of an exemplary variable attenuator 101Athat may be used as the variable attenuator 101. The variable attenuator101A includes NPN bipolar junction transistors (BJTs) Q1-Q4, PNP BJTsQ5-Q8, a resistor R1 and two constant current supplies or sinks 201,203. The current sink 201 draws a current I_(A1) and the current sink203 draws a current I_(A2). The emitters of the transistors Q5-Q8 arecoupled to the V_(Supply) signal. The bases of the transistors Q5 and Q7are coupled together and the base of the transistor Q5 is coupled to itscollector. In a similar manner, the bases of the transistors Q6 and Q8are coupled together and the base of the transistor Q6 is coupled to itscollector.

The collectors of the transistors Q5, Q7, Q8 and Q6 are coupled to thecollectors of the transistors Q1, Q3, Q4 and Q2, respectively. Theemitters of the transistors Q1 and Q3 are coupled together and to oneend of the current sink 201, having its other end coupled to ground. Theemitters of the transistors Q2 and Q4 are coupled together and to oneend of the current sink 203, having its other end coupled to ground. Thebases of the transistors Q1 and Q2 received the V_(IN+) and V_(IN−)signals, respectively, which are the same as the V_(A) and V_(B)signals, respectively. The bases of the transistors Q3 and Q4 developthe V_(C) and V_(D) signals, respectively. The resistor R1 is has oneend coupled to the base of the transistor Q3 and its other end coupledto the base of the transistor Q4. A current I_(D) is defined as flowingfrom the collector to the base of the transistor Q3 and is the samecurrent flowing from the base to the collector of the transistor Q4. Acurrent I_(R1) is defined as flowing through the resistor R1 between theV_(C) and V_(D) signals.

The four output voltages V_(A)-V_(D) are a function of the differentialinput voltage V_(IN) with the V_(A) and V_(B) signals being the same asthe two V_(IN+) and V_(IN−) input signals. There are two differentialpairs of transistors, Q1, Q3 and Q2, Q4, connected to each end of theresistor R1. In the embodiment shown, the transistors Q1-Q4 aresubstantially identical. The transistors Q3 and Q4 are diode connected,having their bases and collectors shorted together. The transistors Q5and Q7 are configured as a first current mirror 205 and the transistorsQ6 and Q8 are configured as a second current mirror 207. In this manner,the collectors of the transistors Q1 and Q2 are connected to the inputsof the current mirrors 205 and 207 with the current mirror outputsconnected to the collectors of the transistors Q3 and Q4. The currentmirrors 205 and 207 are shown as PNP transistors but may be implementedwith any other current mirror configuration. For example, the currentmirrors 205 and 207 may be implemented using other types of transistors,such as PMOS transistors or the like. Also, in the embodiment shown thecurrent mirrors 205 and 207 have an output/input ratio of 1:1, thecurrent sources 201 and 203 are assumed to be matched, and the basecurrents are assumed to be negligible.

The combination of a differential pair of transistors Q1 and Q3 with thefirst current mirror 205 forms a differential- to single-endedtransconductance stage with an output current (I_(D)) equal to thedifferential current out of the differential pair. This current alsoflows through the resistor R1 (I_(R1)), and is defined according to thefollowing equation 2:

I _(D) =I _(R1) =I _(A) tan h[(V _(A) −V _(C))/2V _(T) ]=I _(A) tan h[(V_(D) −V _(B))/2V _(T)],  (EQ 2)

where I_(A2)=I_(A1)=I_(A). The differential input voltage is equal tothe current-resistive (IR) drop across the resistor R1 plus the twodifferential input voltages of the two differential pairs as illustratedin the following equations 3 and 4:

V _(IN+) −V _(IN−)=(V _(A) −V _(C))+(I _(D))R 1+(V _(D) −V _(B))=2(V_(A) −V _(C))+(I _(A))(R 1)tan h[(V _(A) −V _(C))/2V _(T)]  (EQ 3)

V _(IN+) −V _(IN−)˜2(V _(A) −V _(C))+[(I _(A))(R 1)/2V _(T)](V _(A) −V_(C))=(V _(A) −V _(C))(2+[(I _(A))(R 1)/2V _(T)])  (EQ 4)

where □(V_(A)−V_(C))□<V_(T). The following equation 5 is the attenuatortransfer function showing that the voltage applied to eitherdifferential pair is a fraction less than or equal to one-half of theinput voltage:

(V _(A) −V _(C))=(V _(IN+) −V _(IN−))(2V _(T)/[4V _(T)+(I _(A))(R1)])  (EQ 5)

where, again, □(V_(A)−V_(C))□<V_(T). This fraction becomes smaller asthe bias current I_(A) gets larger. This fraction is independent oftemperature when I_(A) is made Proportional To Absolute Temperature(PTAT). The following equation 6 shows the conditions for linearoperation:

□(V _(A) −V _(C))□<V _(T)□□(V _(IN+) −V _(IN−))□<(2V _(T)+(I _(A))(R1)/2)  (EQ 6)

The maximum input voltage may be increased if the current I_(A) isincreased. Increasing I_(A) also increases the amount of attenuation.

FIG. 3 is a schematic diagram of an exemplary transconductance stage103A that may be used as the transconductance stage 103. Thetransconductance stage 103A includes four NPN BJTs Q11, Q12, Q13 and Q14and two current sinks 301 and 303. The transistors Q11 and Q13 arecoupled together at their emitters and to one end of the current sink301 that sinks a current of I_(B1). The transistors Q12 and Q14 arecoupled together at their emitters and to one end of the current sink303 that sinks a current of I_(B2). The current sinks 301, 303 are alsocoupled to ground. The V_(A)-V_(D) signals are provided to the bases ofthe transistors Q11-Q14, respectively. The collectors of the transistorsQ11 and Q14 are coupled together and develop the I_(OUT+) output signal.The collectors of the transistors Q12 and Q13 are coupled together anddevelop the I_(OUT−) output signal. Since the four outputs from thevariable attenuator 101 are the four input signals to thetransconductance stage (V_(A), V_(B), V_(C) and V_(D)), the followingequation 7 describes the differential output current I_(OUT):

(I _(OUT+) −I _(OUT−))=I _(B1) tan h[(V _(A) −V _(C))/2V _(T) ]+I _(B2)tan h[(V _(D) −V _(B))/2V _(T)]=2I _(B) tan h[(V _(A) −V _(C))/2V_(T)]  (EQ 7)

where I_(B2)=I_(B1)=I_(B). Substituting the attenuator transfer function(EQ 5) into the transconductance function above, yields the followingequation 8:

(I _(OUT+) −I _(OUT−))=2(I _(B) /I _(A))I _(R1)=(I _(B) /V _(T))(V_(A)−V_(C))=(V _(IN+) −V _(IN−))(2(I _(B))/[4V _(T)+(I _(A))(R 1])  (EQ8)

where □(V_(A)−V_(C))□<V_(T). The transconductance g_(m) of thetransconductance amplifier 100 employing the variable attenuator 101Aand the transconductance stage 103A is defined according to thefollowing equation 9:

g _(m)=(I _(OUT+) −I _(OUT−))/(V _(IN+) −V _(IN−))=2(I _(B))/[4V _(T)+(I_(A))(R 1)]  (EQ 9)

The differential output current is proportional to the current I_(R1)flowing through the resistor R1. The transconductance can be varied bychanging either the bias current to the transconductance stage (I_(B))or the bias current to the variable attenuator stage (I_(A)) or byvarying both.

FIG. 4 is a schematic diagram of another exemplary variable attenuator101B that may be used as the variable attenuator 101. The variableattenuator 101B is similar to the variable attenuator 101A except thatthe transistors Q5-Q7 are removed. The current sinks 201 and 203 areshown as both providing a current I_(A) (where I_(A) may be equal toI_(A1) and I_(A2)). Also, the collectors of the transistors Q1 and Q2are coupled to the V_(Supply) signal. Finally, two current sources 401and 403 are added, each having one end coupled to the V_(Supply) signaland another end coupled to a respective collector of the transistors Q3and Q4. Each of the current sources 401 and 403 source a current ofI_(A)/2 into the respective collectors of the transistors Q3 and Q4. Itis noted that for high frequency applications, PNP or PMOS currentmirrors may not have the signal bandwidth, so that current sources maybe used as shown in FIG. 4.

The combination of the differential pair of transistors Q1 and Q3 withthe current source 401 forms a differential- to single-endedtransconductance stage, with an output current (I_(D)) equal to half ofthe differential current out of the differential pair. The current I_(D)also flows through the resistor R1 as illustrated by the followingequation 10:

I _(D) =I _(R1)=(I _(A)/2)tan h[(V _(A) −V _(C))/2V _(T)]=(I _(A)/2)tanh[(V _(D) −V _(B))/2V _(T)]  (EQ 10)

The differential input voltage V_(IN) is equal to the IR drop across theresistor R1 plus the two differential input voltages of the twodifferential pairs as illustrated by the following equations 11 and 12:

V _(IN+) −V _(IN−)=(V _(A) −V _(C))+(I _(D))R 1+(V _(D) −V _(B))=2(V_(A) −V _(C))+(I _(A)/2)(R 1)tan h[(V _(A) −V _(C))/2V _(T)]  (EQ 11)

V _(IN+) −V _(IN−)=2(V _(A) −V _(C))+[(I _(A))(R 1)/4V _(T)](V _(A) −V_(C))=(V _(A) −V _(C))(2+[(I _(A))(R 1)/4V _(T)])  (EQ 12)

where □(V_(A)−V_(C))□<V_(T). The voltage differential V_(A)−V_(C) isdefined according to the following equation 13:

(V _(A) −V _(C))=(V _(IN+) −V _(IN−))(4V _(T)/[8V _(T)+(I _(A))(R1)])  (EQ 13)

where □(V_(A)−V_(C))□<V_(T). Thus, the current passing through theresistor R1 is half as large, so that the IR voltage drop is also half.

It is noted that the above discussion assumes that the current sources401, 403 at the collectors of Q3 and Q4 track the current sinks 201, 203at the emitters. These currents I_(A) and I_(A)/2 are varied togetherfor the electronic gain control. One practical way to do this is to usethe current mirror arrangement of the variable attenuator 101A, withcurrent mirrors that do not respond to the signal, i.e., the currentmirror frequency response is much lower than that of the signal.

FIG. 5 is a schematic diagram of another exemplary variable attenuator101C that may be used as the variable attenuator 101. Under somecircumstances, the collector currents of the transistors Q1 and Q2 canbe used without current mirrors. The variable attenuator 101C is similarto the variable attenuator 101B with two additional current sources 501and 503, each coupled between the V_(Supply) signal and the collector ofa respective one of the transistors Q1 and Q2, and each sourcing acurrent of I_(A)/2. Also, a cross-coupled connection is made where thecollector of the transistor Q1 is coupled to the collector of thetransistor Q4 and the collector of the transistor Q2 is coupled to thecollector of the transistor Q3. A current equal to I_(D) flows from thecollector of the transistor Q2 to the collector of the transistor Q3.Also, another current equal to I_(D) flows from the collector of thetransistor Q4 to the collector of the transistor Q1.

The combination of the differential pair of transistors Q1 and Q3 withthe current sources 401 and 501 forms a differential- to single-endedtransconductance stage with two output currents (I_(D)), each equal tohalf of the differential current out of the differential pair. Theseoutput currents are 180 degrees out of phase and are connected toopposite ends of the resistor R1. The current through the resistor R1for the variable attenuator 101C is defined by the following equation14:

I _(R1)=2I _(D) =I _(A) tan h[(V _(A) −V _(C))/2V _(T) ]=I _(A) tan h[(V_(D) −V _(B))/2V _(T)]  (EQ 14)

The differential input voltage is equal to the IR drop across theresistor plus the two differential input voltages of the twodifferential pairs as illustrated by the following equations 15 and 16:

V _(IN+) −V _(IN−)=(V _(A) −V _(C))+(I _(D))R 1+(V _(D) −V _(B))=2(V_(A) −V _(C))+(I _(A))(R 1)tan h[(V _(A) −V _(C))/2V _(T)]  (EQ 15)

V _(IN+) −V _(IN−)=2(V _(A) −V _(C))+[(I _(A))(R 1)/2V _(T)](V _(A) −V_(C))=(V _(A) −V _(C))(2+(I _(A))(R 1)/2V _(T))  (EQ 16)

where □(V_(A)−V_(C))□<V_(T). Solving for V_(A)−V_(C) results in thefollowing equation 17:

 (V _(A) −V _(C))=(V _(IN+) −V _(IN−))(2V _(T)/[4V _(T)+(I _(A))(R1)])  (EQ 17)

where □(V_(A)−V_(C))□<V_(T). The current passing through the resistor R1is the same as the current mirror version, or the variable attenuator101A. The voltage drop across the resistor R1 appears across thebase-collector junctions of the transistors Q1 and Q2. If the variableattenuator 101C is required to handle large input voltages, thetransistors Q1 and Q2 could be driven into saturation unless appropriatelevel shifting elements are added. For example, in an alternativeembodiment, diodes (not shown) are connected in series with the bases ofthe transistors Q3 and Q4. For high frequency operation, the collectorsof the transistors Q1 and Q2 may be coupled to the resistor R1 throughcapacitors (not shown).

FIG. 6 is a schematic diagram of another exemplary variable attenuator101D that may be used as the variable attenuator 101. For high frequencyoperation, the collectors of the transistors Q1 and Q2 are coupled tothe resistor R1 through capacitors C1 and C2, which are used inconjunction with a slow current mirror circuit. The variable attenuator101D is similar to the variable attenuator 101A, except that a capacitorC1 is coupled between the base of the transistor Q8 and the collector ofthe transistor Q3. Also, a capacitor C2 is coupled between the base ofthe transistor Q7 and the collector of the transistor Q4. The currentsinks 201, 203 are shown as sinking currents I_(A1) and I_(A2),respectively.

At low frequencies, the collector currents of the transistors Q1 and Q2are mirrored by the current mirrors 205 and 207, respectively, and aresubtracted from the collector currents of the transistors Q3 and Q4,respectively. At high frequencies, the collector currents of thetransistors Q1 and Q2 go through the capacitors C2 and C1, respectively,and are added to the collector currents from the transistors Q4 and Q3,respectively. The current through the resistor R1 does not change withfrequency.

There are many other variations that are possible and contemplated forthe variable attenuator 101. The resistor R1 may instead be an inductor,a capacitor or some other combination of components as appropriate forthe specific application and signal frequency. The transistors Q1 and Q3may have the same or different emitter areas. Current mirrors, if used,may have a current mirror ratio different than 1:1. The transistorsQ1-Q4 may be replaced by other types of transistor devices, such asmetal oxide semiconductor field effect transistors (MOSFETs) or the likeor other active devices. Also, PNPs may be substituted for NPNs andvice-versa.

FIG. 7 is a schematic diagram of another exemplary transconductancestage 103B that may be used as the transconductance stage 103. Thetransconductance stage 103B is very similar to the transconductancestage 103A, except that the emitters of the transistors Q11-Q14 are allcoupled together and the current sinks 301 and 303 each sink a currentof I_(B) (where I_(B) may be equal to I_(B1) and I_(B2)).

The output current of the transconductance stage 103B is substantiallyidentical to that of the transconductance stage 103A. The two currentsources 301 and 303 may be combined into a single source of twice thevalue. For an input differential voltage of zero, so thatV_(A)=V_(B)=V_(C)=V_(D), the current divides equally between the fourtransistors Q11-Q14 and the differential output current I_(OUT) is alsozero. For a large positive input voltage, the transistors Q12 and Q14are cut off so that the differential current is provided by thefollowing equation 18:

(I _(OUT+) −I _(OUT−))=2(I _(B))tan h[(V _(A) −V _(C))/2V _(T)]  (EQ 18)

where V_(C)−V_(D)>2V_(T). For large negative voltages, the transistorsQ11 and Q13 are cut off. Redefining the currents I_(B1) and I_(B2) asthe currents flowing into the emitters of the differential pairs oftransistors Q11, Q13 and Q12, Q14, respectively, then the differentialoutput current is provided by the following equation 19:

(I _(OUT+) −I _(OUT−))=I _(B1) tan h[(V _(A) −V _(C))/2V _(T) ]+I _(B2)tan h[(V _(D) −V _(B))/2V _(T)]=2(I _(B))tan h[(V _(A) −V _(C))/2V_(T)]  (EQ 19)

where I_(B2)=I_(B1)=2I_(B), and V_(A)−V_(C)=V_(D)−V_(A), which is thesame as EQ 7. If the circuit works the same with either an open or shortbetween the pairs of emitters (the two current sources), then it workswith any two terminal elements connecting the pairs of emitters.

There are also many other variations that are possible and contemplatedfor the transconductance stage 103. For example, the emitter areas ofthe transistors Q11 and Q13 may be equal, although they do not have tobe equal. The transistors Q11-Q14 may be replaced by other types oftransistors, such as MOSFETs or the like, or by other active devices.Also, PNPs may be substituted for NPNs.

The variable transconductance amplifier described herein provide manybenefits and advantages over transconductance amplifiers previouslyavailable. The attenuator, and the entire amplifier, has a relativelyhigh input impedance so that it may easily be driven by a low impedancesource. The attenuator embodiments are each variable in that the gain isvariable and may be electronically controlled. For example, the currentsinks 201 and 203 defining the bias current may each be readilyimplemented in an electronically controllable manner so that theattenuator gain may be variable and controllable as desired. Increasingthe bias current increases the maximum input voltage of the differentialinput voltage V_(IN) while still maintaining linearity. Increasing thebias current also increases the amount of attenuation. The current sinks301 and 303 defining the bias current of the transconductance stage 103may also be implemented to be electronically controlled in a similarmanner to vary the gain and thus the transconductance of the overallvariable transconductance amplifier 100. Of course, the bias current ofwither or both of the variable attenuator and transconductance stagesmay be controlled to adjust gain and transconductance as desired. Inthis manner, the maximum useful input voltage may be increased whilemaintaining linearity.

Although a system and method according to the present invention has beendescribed in connection with one or more embodiments, it is not intendedto be limited to the specific form set forth herein, but on thecontrary, it is intended to cover such alternatives, modifications, andequivalents, as can be reasonably included within the spirit and scopeof the present invention as defined by the appended claims.

What is claimed is:
 1. A variable transconductance amplifier,comprising: a variable attenuator, comprising: first and seconddifferential to single-ended transconductance stages each biased by arespective one of first and second current devices, and each having afirst control terminal for collectively receiving a differential inputvoltage signal, and each having a second control terminal that developsa current signal; and at least one reactive element coupled between thesecond control terminals of the first and second differential tosingle-ended transconductance stages; and a transconductance stage,comprising: a first differential pair having first and second controlterminals and first and second output terminals, wherein the first andsecond control terminals of the first differential pair are coupled tothe first and second control terminals, respectively, of the firstdifferential to single-ended transconductance stage of the variableattenuator; a second differential pair having first and second controlterminals and first and second output terminals, wherein the first andsecond control terminals of the second differential pair are coupled tothe first and second terminals, respectively, of the second differentialto single-ended transconductance stage of the variable attenuator; andwherein the first output terminal of the first differential pair iscoupled to the second output terminal of the second differential pairand forms a first polarity of a differential output current signal andwherein the first output terminal of the second differential pair iscoupled to the second output terminal of the first differential pair andforms a second polarity of the differential output current signal. 2.The variable transconductance amplifier of claim 1, wherein the at leastone reactive element comprises a resistor.
 3. The variabletransconductance amplifier of claim 1, wherein the at least one reactiveelement comprises a combination of elements having a predeterminedfrequency response.
 4. The variable transconductance amplifier of claim1, wherein the first and second current devices are electronicallycontrollable to adjust the gain of the variable attenuator and to adjustthe maximum allowable input voltage of the differential input voltagewhile maintaining linearity.
 5. The variable transconductance amplifierof claim 1, wherein the first and second differential pairs are eachbiased by at least one current device that is electronicallycontrollable to adjust the transconductance between the differentialinput voltage signal and the differential output current signal.
 6. Thevariable transconductance amplifier of claim 1, wherein the first andsecond differential pairs are biased by third and fourth currentdevices, respectively, and wherein the first, second, third and fourthcurrent devices are controllable to adjust the overall transconductancebetween the differential input voltage signal and the differentialoutput current signal.
 7. The variable transconductance amplifier ofclaim 1, wherein the first and second differential pairs of thetransconductor stage each have a bias terminal, and wherein the biasterminals of the first and second differential pairs are coupledtogether.
 8. The variable transconductance amplifier of claim 7, furthercomprising: a third current device coupled to the common bias terminalsof the first and second differential pairs.
 9. The variabletransconductance amplifier of claim 1, wherein the first and seconddifferential to single-ended transconductance stages of the variableattenuator each comprise: a differential pair of transistors including adiode-coupled transistor; a respective one of the first and secondcurrent devices comprising a current sink coupled to the differentialpair; and a current mirror coupled to the differential pair.
 10. Thevariable transconductance amplifier of claim 1, wherein the first andsecond differential to single-ended transconductance stages of thevariable attenuator each comprise: a differential pair of transistorsincluding a first transistor coupled to a voltage supply signal and adiode-coupled transistor; a respective one of the first and secondcurrent devices comprising a current sink coupled to the differentialpair; and a current source coupled to the diode-coupled transistor ofthe differential pair, the current source sourcing a current that isproportional to the current developed by a corresponding current sink.11. The variable transconductance amplifier of claim 1, wherein thefirst and second differential to single-ended transconductance stages ofthe variable attenuator each comprise: a differential pair oftransistors including a diode-coupled transistor; a respective one ofthe first and second current devices comprising a current sink coupledto the differential pair; first and second current sources, each coupledto source current to a respective one of the transistors of thedifferential pair, the current sources each sourcing a current that isproportional to the current developed by a corresponding one of thefirst and second current sinks.
 12. The variable transconductanceamplifier of claim 11, wherein the differential pairs of transistors ofthe first and second differential to single-ended transconductancestages are cross-coupled.
 13. A variable transconductance amplifier,comprising: a variable attenuator, comprising: a first differential tosingle-ended transconductance stage, biased by a first current deviceand having first and second control terminals, that develops a currentsignal via the second control terminal; a second differential tosingle-ended transconductance stage, biased by a second current deviceand having first and second control terminals, that develops a currentsignal via the second control terminal; and a reactive circuit coupledbetween the second control terminal of the first differential tosingle-ended transconductance stage and the second control terminal ofthe second differential to single-ended transconductance stage; whereinthe first control terminal of each of the first and second differentialto single-ended transconductance stages receive a differential inputvoltage signal; and a transconductance stage, comprising: a firstdifferential pair, biased by a third current device and having first andsecond control terminals and first and second output terminals, whereinthe first and second control terminals of the first differential pairare coupled to the first and second control terminals, respectively, ofthe first differential to single-ended transconductance stage of thevariable attenuator; a second differential pair, biased by a fourthcurrent device and having first and second control terminals and firstand second output terminals, wherein the first and second controlterminals of the second differential pair are coupled to the first andsecond terminals, respectively, of the second differential tosingle-ended transconductance stage of the variable attenuator; andwherein the first output terminal of the first differential pair iscoupled to the second output terminal of the second differential pairand forms a first polarity of a differential output current signal andwherein the first output terminal of the second differential pair iscoupled to the second output terminal of the first differential pair andforms a second polarity of the differential output current signal. 14.The variable transconductance amplifier of claim 13, wherein thevariable attenuator comprises: a first current mirror, coupled to avoltage supply signal, having an input and an output; a second currentmirror, coupled to the voltage supply signal, having an input and anoutput; the first and second current devices each comprising a currentsink referenced to a common reference signal; a first bipolar junctiontransistor having a base receiving a first polarity of the differentialinput voltage signal, a collector coupled to the input of the firstcurrent mirror and an emitter coupled to the first current sink; asecond bipolar junction transistor having a base receiving a secondpolarity of the differential input voltage signal, a collector coupledto the input of the second current mirror and an emitter coupled to thesecond current sink; and a third bipolar junction transistor having abase coupled to a first end of the reactive element, a collector coupledto its base and to the output of the first current mirror, and anemitter coupled to the emitter of the first transistor; a fourth bipolarjunction transistor having a base coupled to a second end of thereactive element, a collector coupled to its base and to the output ofthe second current mirror, and an emitter coupled to the emitter of thesecond transistor.
 15. The variable transconductance amplifier of claim14, wherein the first, second, third and fourth transistors are NPNbipolar junction transistors.
 16. The variable transconductanceamplifier of claim 14, wherein the first and second current sinks areelectronically controllable.
 17. The variable transconductance amplifierof claim 14, wherein the first and second current mirrors each comprise:a first PNP bipolar junction transistor having an emitter coupled to thevoltage supply signal, and a base and a collector coupled together andforming an input terminal of the respective current mirror; and a secondPNP bipolar transistor having an emitter coupled to the voltage supplysignal, a base coupled to the base of the first PNP transistor and acollector forming an output of the respective current mirror.
 18. Thevariable transconductance amplifier of claim 17, further comprising: afirst capacitor coupled between the collector of a second PNP transistorof the first current mirror and to the common bases of the first andsecond PNP transistors of the second current mirror; and a secondcapacitor coupled between the collector of a second PNP transistor ofthe second current mirror and to the common bases of the first andsecond PNP transistors of the first current mirror.
 19. The variabletransconductance amplifier of claim 13, wherein the reactive circuitcomprises a resistor.
 20. The variable transconductance amplifier ofclaim 13, wherein the reactive circuit comprises at least one elementcoupled to provide a predetermined frequency response.
 21. The variabletransconductance amplifier of claim 13, wherein the variable attenuatorcomprises: a first current source having an input coupled to a voltagesupply signal and an output; a second current source having an inputcoupled to a voltage supply signal and an output; the first and secondcurrent devices each comprising a current sink referenced to a commonreference signal; a first bipolar junction transistor having a basereceiving a first polarity of the differential input voltage signal, acollector coupled to the voltage supply signal and an emitter coupled tothe first current sink; a second bipolar junction transistor having abase receiving a second polarity of the differential input voltagesignal, a collector coupled to the voltage supply signal and an emittercoupled to the second current sink; a third bipolar junction transistorhaving a base coupled to a first end of the reactive element, acollector coupled to its base and to the output of the first currentsource, and an emitter coupled to the emitter of the first transistor; afourth bipolar junction transistor having a base coupled to a second endof the reactive element, a collector coupled to its base and to theoutput of the second current source, and an emitter coupled to theemitter of the second transistor.
 22. The variable transconductanceamplifier of claim 21, wherein the first and second current sinks eachsink approximately the same amount of current as each other and whereinthe first and second current sources each source approximately the sameamount of current as each other.
 23. The variable transconductanceamplifier of claim 22, wherein the first and second current sources eachsource approximately half the current of the first and second currentsinks.
 24. The variable transconductance amplifier of claim 13, whereinthe variable attenuator comprises: first, second, third and fourthcurrent sources, each having an input coupled to a voltage supply signaland an output; the first and second current devices each comprising acurrent sink referenced to a common reference signal; a first bipolarjunction transistor having a base receiving a first polarity of thedifferential input voltage signal, a collector coupled to the output ofthe first current source and an emitter coupled to the first currentsink; a second bipolar junction transistor having a base receiving asecond polarity of the differential input voltage signal, a collectorcoupled to the output of the fourth current source and an emittercoupled to the second current sink; a third bipolar junction transistorhaving a base coupled to a first end of the reactive element, acollector coupled to its base and to the outputs of each of the secondand fourth current sources, and an emitter coupled to the emitter of thefirst transistor; a fourth bipolar junction transistor having a basecoupled to a second end of the reactive element, a collector coupled toits base and to the outputs of each of the third and fourth currentsources, and an emitter coupled to the emitter of the second transistor.25. The variable transconductance amplifier of claim 24, wherein thefirst and second current sinks each sink approximately the same amountof current as each other and wherein the first, second, third and fourthcurrent sources each source approximately the same amount of current aseach other.
 26. The variable transconductance amplifier of claim 25,wherein the first, second, third and fourth current sources each sourceapproximately half the current of the first and second current sinks.27. The variable transconductance amplifier of claim 13, wherein thetransconductance stage comprises: the third and fourth current deviceseach comprising a current sink referenced to a common reference signal;a first bipolar junction transistor having an emitter coupled to thethird current sink, a base receiving the first polarity of thedifferential input voltage signal and a collector that develops a firstpolarity of the differential output voltage signal; and a second bipolartransistor having an emitter coupled to the emitter of the firsttransistor, a base forming the second control terminal of the firstdifferential pair and a collector; a third bipolar junction transistorhaving an emitter coupled to the fourth current sink, a base receivingthe second polarity of the differential input voltage signal and acollector coupled to the collector of the second transistor fordeveloping a second polarity of the differential output voltage signal;and a fourth bipolar junction transistor having an emitter coupled tothe emitter of the third transistor, a base forming the second controlterminal of the second differential pair and a collector and a collectorcoupled to the collector of the first transistor for developing thefirst polarity of the differential output voltage signal.
 28. Thevariable transconductance amplifier of claim 27, wherein the first,second, third and fourth transistors are NPN bipolar junctiontransistors.
 29. The variable transconductance amplifier of claim 27,wherein the emitters of the first, second, third and fourth transistorsare coupled together.
 30. The variable transconductance amplifier ofclaim 29, wherein the third and fourth current sinks comprise a single,electronically controllable current sink.
 31. A variabletransconductance amplifier, comprising: a transconductance stageincluding first and second differential pairs each havingcurrent-controlled differential outputs that are cross-coupled togetherto form a differential current output, the first and second differentialpairs each having first and second inputs; and a variable attenuatorincluding first and second differential to single-ended transconductancestages, each having a first terminal for collectively receiving adifferential input voltage signal and a second terminal, wherein thesecond terminals are coupled together through a reactive element,wherein the first and second terminals of the first differential tosingle-ended transconductance stage are coupled to the first and secondinputs of the first differential pair, respectively, and wherein thefirst and second terminals of the second differential to single-endedtransconductance stage are coupled to the first and second inputs of thesecond differential pair, respectively.
 32. The variabletransconductance amplifier of claim 31, wherein the first and seconddifferential to single-ended transconductance stages of the variableattenuator are biased by at least one electronically controllablecurrent device.
 33. The variable transconductance amplifier of claim 31,wherein the first and second differential pairs of the transconductancestage are biased by at least one electronically controllable currentdevice.